The present invention generally relates to a full duplex transceiver in portable cellular phone systems, pager systems, etc., and more specifically to an error estimator which compensates for various disturbances caused to signals being received in a direct conversion receiver located within the transceiver.
Receivers in cellular systems and other fields noted above are preferably small, lightweight and inexpensive. To make a portable receiver such as a hand-held telephone smaller and less expensive, the integration of parts has become very important. Heterodyne receivers usually are of high cost to produce and have many parts such as bandpass filters that are unable to be integrated. To overcome such drawbacks, direct conversion receiver architecture has been developed in which the frequency of the local oscillator is the same as the frequency of the received radio carrier. Consequently, the received radio signal is down-converted directly to base band in one step. Since a direct-conversion receiver does not have any intermediate frequency (IF) stages, many filters can be omitted or simplified.
Direct conversion was introduced for single-sideband receivers in the 1950""s, but the technique is not limited to such systems. Direct conversion can be used with many different modulation schemes and is especially well suited for the quadrature modulation schemes of today, such as minimum shift keying (MSK) and quadrature amplitude modulation (QAM). Various aspects of direct-conversion receivers are described in U.S. Pat. No. 5,530,929 entitled xe2x80x9cRadio Receiver.xe2x80x9d
The operation of a conventional direct-conversion receiver can be described as follows with reference to FIG. 1. A radio frequency (RF) signal having center frequency fc and bandwidth BWrf is received by an antenna 10 and then is filtered by a bandpass filter 20. The filtered signal produced by the bandpass filter is amplified by an amplifier 30, which preferably has low noise to improve the total noise figure of the receiver.
The amplified and filtered signal produced by the amplifier 30 is then down-converted to base band in an in-phase (I) channel and a quadrature phase (Q) channel by balanced mixers 40, 50. The mixers are driven by respective ones of sine (I) and cosine (Q) signals, produced from a sinusoidal signal generated by a local oscillator 60, by a suitable divider and phase shifter 70. According to the direct-conversion principle, the local oscillator signal also has the frequency fc.
The mixers 40, 50 effectively multiply the signal from the amplifier 30 and the I and Q signals of the local oscillator. Each mixer produces a signal that has frequencies that are the sum and difference of the frequencies of the amplified filtered received signal and the local oscillator signal. The difference (down-converted) signals each have a spectrum that is folded over around zero frequency (DC) and that spans from DC to xc2xdBWrf.
The I and Q signals produced by the mixers 40, 50 are filtered by low-pass filters 80, 90 that remove the sum (up-converted) signals, as well as components that might be due to nearby RF signals. The filters 80, 90 set the noise bandwidth, and thus, the total noise power in the receiver. The I and Q signals are then amplified by amplifiers 100, 110, and provided to further processing components that produce the demodulated output signal. The further processing can include phase demodulation, amplitude demodulation, frequency demodulation, or hybrid demodulation schemes.
One major problem with the direct-conversion receiver is that baseband signal distortion can be caused by a pure DC signal generated by local oscillator leakage, in addition to second-order products of interferers (e.g., signals on the same and nearby RF communication channels) produced by active elements located close to the receiver. The distortions, located at the base band, interfere with the desired base band signal, thereby degrading performance of a direct conversion receiver. In some situations, this problem totally blocks communication in high-performance receivers for today""s time division multiple access (TDMA) and wide band code division multiple access (WCDMA) digital cellular systems.
A second order non-linearity together with a strong constant envelope RF interferer will cause a DC component within a received signal. Such a DC component can be blocked, for example, with a DC blocking capacitor.
Amplitude Modulated (AM) interferers, however, present a larger problem since the disturbance at baseband will not be a pure DC signal and therefore cannot be easily removed. In non-linear devices such as an amplifier, an input signal, Vin, will produce an output signal, Vout. The characteristics of an amplifier located within a transceiver can be defined as Vout=c1Vin+c2Vin2+c3Vin3+ . . . , where an input signal can be Vin(t)=v1(t)xc2x7Cos(xcfx891t). If the input signal is applied to the input of the amplifier, the output is described by the following equation:                                           v            out                    ⁡                      (            t            )                          =                                            1              2                        ⁢                          c              2                        ⁢                                          v                1                2                            ⁡                              (                t                )                                              +                                    (                                                                    c                    1                                    ⁢                                                            v                      1                                        ⁡                                          (                      t                      )                                                                      +                                                      3                    4                                    ⁢                                      c                    3                                    ⁢                                                            v                      1                      3                                        ⁡                                          (                      t                      )                                                                                  )                        ⁢            cos            ⁢                          xe2x80x83                        ⁢                          ω              1                        ⁢            t                    +                                    1              2                        ⁢                          c              2                        ⁢                                          v                1                2                            ⁡                              (                t                )                                      ⁢            cos            ⁢                          xe2x80x83                        ⁢            2            ⁢                          ω              1                        ⁢            t                    +                                    1              4                        ⁢                          c              3                        ⁢                          v              1              3                        ⁢            cos            ⁢                          xe2x80x83                        ⁢            3            ⁢                          ω              1                        ⁢            t                    +          …                                    (        1        )            
From this equation it can be seen that the input signal will generate a baseband distortion component described by:                                                         v              bb                        ⁡                          (              t              )                                =                                    1              2                        ⁢                          c              2                        ⁢                                          v                1                2                            ⁡                              (                t                )                                                    ,                            (        2        )            
where constant c2 depends on a second order intercept point of the amplifier. The second order intercept point is determined by applying two signals at two different frequencies, f1 and f2, to the amplifier. The output power, Pout, is plotted at the first frequency, the sum of the first and second frequency against the input power Pin of either f1 or f2. Extrapolation of Pout versus Pin of either f1 or f2 yields the second order intercept point. If ViIP2 is the voltage input of the amplifier at the second order intercept point, the baseband voltage can be written as:                                           v            bb                    ⁡                      (            t            )                          =                              1            2                    ⁢                                    c              1                                      V              iIP2                                ⁢                                                    v                1                2                            ⁡                              (                t                )                                      .                                              (        3        )            
In an application which consists of both I and Q modulators the interference problem can be described with reference to FIG. 2. An interfering signal, IF, is an AM modulated signal that generates baseband interference in both the Q and I channels, where
VbbI(t)=Kixc2x7v12(t)
VbbQ(t)=Kqxc2x7v12(t),xe2x80x83xe2x80x83(4)
and Ki and Kq are constants. The disturbance on the I and Q channels, however, is not necessarily equal, due to the differing levels of interfering signals on the channels, and the different components in the IQ demodulator. The resulting signal is an error signal that is combined with the desired signal. The error signal can be written as:
xe2x80x83xcex5i(t)=Kixc2x7v12(t)
xcex5q(t)=Kqxc2x7v12(t),xe2x80x83xe2x80x83(5)
The AM interferer together with second order non-linearities in the receiver generate a baseband error vector. The phase of the error vector is constant, or varies slightly, and the magnitude is proportional to the squared envelope of the interferer. Since the disturbance is not necessarily equal on I and Q channels, Ki is not necessarily equal to Kq in the general case. Accordingly, the error signal can be written as:
xcex5(t)=xcex5i(t)+jxcex5q(t)=rxcex5(t)ejy,xe2x80x83xe2x80x83(6)
where
rxcex5(t)=Kyxc2x7v12(t)
Ky=|Ki+jKq|
and
xcex4=arg(Ki+jKq)xe2x80x83xe2x80x83(7)
where xcex3 is an arbitrary phase shift that is constant or changes slightly with, for example, a temperature variation.
As discussed above, in the case where the input has a constant envelope, the baseband interference will be a pure DC component. The offset caused by a pure DC component can be compensated for, in the simplest case, by a DC blocking capacitor or as described in U.S. Pat. No. 5,241,702 to Dent. Compensating for a constant RF interference, however, is more complex. For example, if the transceiver is of full duplex type (i.e., transmitting and receiving at the same time) the transmitted signal can be a very strong interferer of the receiver.
Generally, the amplitude of the input signal will be a function of time. The interference will therefore also be a function of time. In U.S. Pat. No. 5,579,347 by Lindquist et al., two methods of removing interference from amplitude modulated (AM) interferers are described.
First, a switched interferer, such as a GSM signal, causes a DC step once every time the power is turned on and off. For a GSM interferer, this occurs once every time slot (e.g., approximately 600 us). The method described by Lindquist removes the DC step caused by a switched interferer. However, the method operates in a system where the interfering DC step occurs once every slot which is a limitation that does not address the general case where the interferer is amplitude modulated.
There have also been several other techniques employed to handle this problem. For example, the transceiver can employ duplex filters. These filters are bulky and have stringent requirements with respect to attenuation of the transmitter in the receive path. The transmitter and receiver can also be separated by large distances and isolated by shielding in order to reduce interference. Additionally, the transceiver can be employed with extremely linear, and therefore current consuming, amplifiers in the receiver.
These techniques reduce the efficiency of the transceiver so much that using direct conversion receivers in full duplex transceiver has not been considered practical. However, systems based on full duplex transceivers are becoming more and more common with the requirement for high bit rates increasing. For example, when utilizing wide band CDMA, such transceivers are in demand. In wide band CDMA; the receiver-transmitter band separation is large (e.g., 130 MHz) which makes the use of a direct conversion receiver more feasible. Therefore, what is needed is the ability to efficiently compensate for disturbances within a direct conversion receiver in a full duplex phone.
To solve the problems associated with disturbance caused by AM interferers and nonlinearities in direct conversion receiver within a transceiver, an error estimator is employed to reduce errors induced on the baseband of the received signal. Knowledge that the transmitter within the transceiver is the strongest interferer to signals being received by that transceiver can be used by the error estimator to subtract the interference caused by transmitter from the received signal. Additionally, this can be achieved even though the interference on the I and Q channels of the receiver is not equal.
In exemplary embodiments of the present invention direct conversion methods and apparatuses are described, including: transmitting signals via the transceiver, receiving input signals at the transceiver, wherein the input signals include desired signals in combination with interfering signals, determining a time delay associated with the interfering signals and correcting, after the time delay, the input signals to compensate for the interfering signals.
In another exemplary embodiment of the present invention correction of interference between a transmitter and receiver, in a transceiver, is described including: a calculation unit for determining a squared envelope signals transmitted by the transmitter, a synchronization unit for determining a time delay associated with reception of the squared envelope by the receiver, a delay unit for applying a time delay to the squared envelope, and an estimator and scaling unit for determining a compensation value to apply to the receiver based on the time delayed, squared envelope.
In yet other exemplary embodiments transceivers used for signal transmission and reception are described which include: a direct conversion receiver for receiving an incoming signal and downconverting the signal into a baseband signal, a transmitter which receives data to be transmitted and modulates the data for transmission to a destination and an error correction device for utilizing the modulated data of the transmitter and a time delay indicating the amount of time necessary for a transmitted signal to interfere with the receiver to compensate the receiver.